View MAX8702_283052.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

19-3357; Rev 0; 8/04
Dual-Phase MOSFET Drivers with Temperature Sensor
General Description
The MAX8702/MAX8703 dual-phase noninverting MOSFET drivers are designed to work with PWM controller ICs, such as the MAX8705/MAX8707, in notebook CPU core and other multiphase regulators. Applications can either step down directly from the battery voltage to create the core voltage, or step down from a low-voltage system supply. The single-stage conversion method allows the highest possible efficiency, while the 2-stage conversion at higher switching frequency provides the minimum possible physical size. Each MOSFET driver is capable of driving 3nF capacitive loads with only 19ns propagation delay and 8ns typical rise and fall times. Larger capacitive loads are allowable but result in longer propagation and transition times. Adaptive dead-time control helps prevent shootthrough currents and maximizes converter efficiency. The MAX8702/MAX8703 feature zero-crossing comparators on each channel. When enabled, these comparators permit the drivers to be used in pulse-skipping operation, thereby saving power at light loads. A separate shutdown control is also included that disables all functions, drops quiescent current to 2A, and sets DH low and DL high. The MAX8702 integrates a resistor-programmable temperature sensor. An open-drain output (DRHOT) signals to the system when the local die temperature exceeds the set temperature. The MAX8702/MAX8703 are available in a thermally-enhanced 20-pin thin QFN package.
Features
o Dual-Phase MOSFET Driver o 0.35 (typ) On-Resistance and 5A (typ) Drive Current o Drives Large Synchronous-Rectifier MOSFETs o Integrated Temperature Sensor (MAX8702 Only) Resistor Programmable Open-Drain Driver Hot Indicator (DRHOT) o Adaptive Dead Time Prevents Shoot-Through o Selectable Pulse-Skipping Mode o 4.5V to 28V Input Voltage Range o Thermally Enhanced Low-Profile Thin QFN Package
MAX8702/MAX8703
Ordering Information
PART TEMP RANGE PINPACKAGE DESCRIPTION
MAX8702ETP -40C to +100C
Dual-Phase 20 Thin QFN Driver with 4mm x 4mm Temp. Sensor Dual-Phase 20 Thin QFN Driver without 4mm x 4mm Temp. Sensor
MAX8703ETP -40C to +100C
Minimal Operating Circuit
+5V VDD VIN 4.5V TO 28V
Applications
Multiphase High-Current Power Supplies 2- to 4-Cell Li+ Battery to CPU Core Supplies Notebook and Desktop Computers Servers and Workstations
+5V
BST1 DH1
VCC AGND
LX1 DL1 PGND1
VOUT
TSET +5V
MAX8702
BST2 DRHOT SHDN LX2 SKIP PWM1 PWM2 DL2 PGND2 DH2 VOUT VIN 4.5V TO 28V
Pin Configuration appears at end of data sheet. ________________________________________________________________ Maxim Integrated Products 1
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
Dual-Phase MOSFET Drivers with Temperature Sensor MAX8702/MAX8703
ABSOLUTE MAXIMUM RATINGS
VCC to AGND............................................................-0.3V to +6V VDD to AGND............................................................-0.3V to +6V PGND_ to AGND ...................................................-0.3V to +0.3V SKIP, SHDN, DRHOT, TSET to AGND......................-0.3V to +6V PWM_ to AGND ........................................................-0.3V to +6V DL_ to PGND_ ............................................-0.3V to (VDD + 0.3V) LX_ to AGND .............................................................-2V to +30V DH_ to LX_ ...............................................-0.3V to (VBST_ + 0.3V) BST_ to LX_ ..............................................................-0.3V to +6V Continuous Power Dissipation (TA = +70C) 20-Pin 4mm x 4mm Thin QFN (derate 16.9mW/C above +70C) .............................1349mW Operating Temperature Range .........................-40C to +100C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 2. VCC = VDD = VSHDN = VSKIP = 5V, TA = 0C to +85C. Typical values are at TA = +25C, unless otherwise noted.)
PARAMETER Input Voltage Range VCC Undervoltage-Lockout Threshold VCC Quiescent Current (Note 1) VDD Quiescent Current VCC Shutdown Current VDD Shutdown Current tPWM-DL tDH-DL tDL-DH tPWM-DH tF_DL tR_DL tF_DH tR_DH RDH RDL_HIGH RDL_LOW IDH IDL_SINK SYMBOL VCC VUVLO ICC IDD 85mV typical hysteresis VCC rising VCC falling CONDITIONS MIN 4.5 3.4 3.3 3.85 3.75 200 2 1 2 1 19 36 25 23 11 8 14 16 1.0 1.0 0.35 1.5 1.5 5 2.5 4.5 4.5 2.0 TYP MAX 5.5 4.1 4.0 400 3 5 5 5 UNITS V V A mA A A A
SKIP = AGND, PWM_ = AGND SKIP = AGND, PWM_ = VCC SKIP = AGND, PWM_ = AGND SHDN = SKIP = AGND SHDN = SKIP = AGND PWM_ high to DL_ low DH_ low to DL_ high DL_ low to DH_ high PWM_ low to DH_ low DL_ falling, 3nF load DL_ rising, 3nF load DH_ falling, 3nF load DH_ rising, 3nF load VBST_ - VLX_ = 5V High state (pullup) Low state (pulldown) VDH_ = 2.5V, VBST_ - VLX_ = 5V VDL_ = 5V VPGND_ - VLX_, SKIP = AGND
GATE DRIVERS AND DEAD-TIME CONTROL (Figure 1) DL_ Propagation Delay DH_ Propagation Delay DL_ Transition Time DH_ Transition Time DH_ On-Resistance (Note 2) DL_ On-Resistance (Note 2) DH_ Source/Sink Current DL_ Source Current DL_ Sink Current Zero-Crossing Threshold TEMPERATURE SENSOR Temperature Threshold Accuracy DRHOT Output Low Voltage DRHOT Leakage Current TA = +85C to +125C, 10C falling hysteresis ISINK = 3mA High state, VDRHOT = 5.5V -5 +5 0.4 1 C V A ns ns ns ns A A A mV
IDL_SOURCE VDL_ = 2.5V
2
_______________________________________________________________________________________
Dual-Phase MOSFET Drivers with Temperature Sensor
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 2. VCC = VDD = VSHDN = VSKIP = 5V, TA = 0C to +85C. Typical values are at TA = +25C, unless otherwise noted.)
PARAMETER Thermal-Shutdown Threshold LOGIC CONTROL SIGNALS Logic Input High Voltage Logic Input Low Voltage Logic Input Current SHDN, SKIP, PWM1, PWM2 SHDN, SKIP, PWM1, PWM2 SHDN, SKIP, PWM1, PWM2 -1 2.4 0.8 +1 V V A SYMBOL 10C hysteresis CONDITIONS MIN TYP +160 MAX UNITS C
MAX8702/MAX8703
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 2. VCC = VDD = VSHDN = VSKIP = 5V, TA = -40C to +100C, unless otherwise noted.) (Note 3)
PARAMETER Input Voltage Range VCC Undervoltage-Lockout Threshold VCC Quiescent Current VDD Quiescent Current VCC Shutdown Current VDD Shutdown Current GATE DRIVERS AND DEAD-TIME CONTROL DH_ On-Resistance (Note 2) DL_ On-Resistance (Note 2) TEMPERATURE SENSOR DRHOT Output Low Voltage LOGIC CONTROL SIGNALS Logic Input High Voltage Logic Input Low Voltage SHDN, SKIP, PWM1, PWM2 SHDN, SKIP, PWM1, PWM2 2.4 0.8 V V ISINK = 3mA 0.4 V RDH RDL _HIGH RDL _LOW VBST_ - VLX_ = 5V High state (pullup) Low state (pulldown) 1.0 1.0 0.35 4.5 4.5 2.0 SYMBOL VCC VUVLO ICC IDD 85mV typical hysteresis VCC rising VCC falling CONDITIONS MIN 4.5 3.4 3.3 TYP MAX 5.5 4.1 4.0 450 3 5 5 5 UNITS V V A mA A A A
SKIP = AGND, PWM_ = PGND_ SKIP = AGND, PWM_ = VCC SKIP = AGND, PWM_ = PGND_, TA = -40C to +85C SHDN = SKIP = AGND, TA = -40C to +85C SHDN = SKIP = AGND, TA = -40C to +85C
Note 1: Static drivers instead of pulsed-level translators. Note 2: Production testing limitations due to package handling require relaxed maximum on-resistance specifications for the thin QFN package. Note 3: Specifications from -40C to +100C are guaranteed by design, not production tested.
_______________________________________________________________________________________
3
Dual-Phase MOSFET Drivers with Temperature Sensor MAX8702/MAX8703
PWM_ 90% 90%
DH_ DL_ tPWM-DL tF_DL 90%
10% tDL-DH tR_DH tPWM-DH tF_DH
10% tDH-DL tR_DL 90%
10%
10%
Figure 1. Timing Definitions Used in the Electrical Characteristics
Typical Operating Characteristics
(Circuit of Figure 2. VIN = 12V, VDD = VCC = VSHDN = VSKIP = 5V, TA = +25C unless otherwise noted.)
POWER DISSIPATION vs. FREQUENCY (SINGLE PHASE, BOTH DRIVERS SWITCHING)
MAX8702/03 toc01
POWER DISSIPATION vs. CAPACITIVE LOAD (SINGLE PHASE, BOTH DRIVERS SWITCHING)
FREQ = 1.2MHz POWER DISSIPATION (mW) 300 FREQ = 0.6MHz 200
MAX8702/03 toc02
350 300 POWER DISSIPATION (mW) 250 200 150 100 50 0 0 0.2 0.4 0.6 0.8 FREQUENCY (MHz) CDL = 3nF, CDH = 1.5nF VCC = 5.5V CDL = 3nF, CDH = 3nF CDL = 6nF, CDH = 3nF
400
100 FREQ = 0.3MHz 0 1 2 3 4 VCC = 5.5V, CDH = CDL 5 6
1.0
1.2
CAPACITANCE (nF)
DL RISE/FALL TIME vs. CAPACITIVE LOAD
MAX8702/03 toc03
DH RISE/FALL TIME vs. CAPACITIVE LOAD
MAX8702/03 toc04
20
30 25 RISE/FALL TIME (ns) DH RISE 20 15 DH FALL 10 5
RISE/FALL TIME (ns)
15
DL RISE
10
DL FALL 5
0 1 2 3 4 5 6 CAPACITANCE (nF)
0 1 2 3 4 5 6 CAPACITANCE (nF)
4
_______________________________________________________________________________________
Dual-Phase MOSFET Drivers with Temperature Sensor MAX8702/MAX8703
Typical Operating Characteristics (continued)
(Circuit of Figure 2. VIN = 12V, VDD = VCC = VSHDN = VSKIP = 5V, TA = +25C unless otherwise noted.)
DH/DL RISE/FALL TIMES vs. TEMPERATURE
DH RISE DH FALL
MAX8702 toc05
PROPAGATION DELAY vs. TEMPERATURE
DL FALL TO DH RISE PROPAGATION DELAY (ns) 40 PWM FALL TO DH FALL 30
MAX8702 toc06
20
50
RISE/FALL TIME (ns)
15
10 DL RISE 5 DL FALL CDH = CDL = 3nF 0 0 20 40 60 80 100 120 TEMPERATURE (C)
20 PWM RISE TO DL FALL 10 CDH = CDL = 3nF 0 0 30 60 90 120 150 TEMPERATURE (C)
RTSET vs. TEMPERATURE
MAX8702 toc07
TYPICAL SWITCHING WAVEFORMS
MAX8702 toc08
70 60 50 RTSET (k)
5V A 0 5V B 0 10V 0 0 125ns/div C. DH, 10V/div D. LX, 10V/div C D
40 30 20 10 0 50 70 90 110 130 150 A. PWM, 5V/div B. DL, 5V/div TEMPERATURE (C)
DH RISE AND DL FALL WAVEFORMS
MAX8702 toc09
DH FALL AND DL RISE WAVEFORMS
MAX8702 toc10
5V A 0 5V B 0 10V 0 D 0 20ns/div C. DH, 10V/div D. LX, 10V/div C
5V A 0 5V 0 10V 0 0 20ns/div C. DH, 10V/div D. LX, 10V/div D B
C
A. PWM, 5V/div B. DL, 5V/div
A. PWM, 5V/div B. DL, 5V/div
_______________________________________________________________________________________
5
Dual-Phase MOSFET Drivers with Temperature Sensor MAX8702/MAX8703
Pin Description
PIN MAX8702 1 2 3 MAX8703 1 2 3 NAME PWM1 PWM2 AGND FUNCTION Phase 1 PWM Logic Input. DH1 is high when PWM1 is high; DL1 is high when PWM1 is low. Phase 2 PWM Logic Input. DH2 is high when PWM2 is high; DL2 is high when PWM2 is low. Analog Ground. The AGND and PGND_ pins must be connected externally at one point close to the IC. Connect the device's exposed backside pad to AGND. Temperature-Set Input. Connect an external 1% resistor from TSET to AGND to set the trip point. RTSET = 85,210 / T - 745,200 / T2 - 195, where RTSET is the temperature-setting resistor in k and T is the trip temperature in Kelvin. Driver-Hot-Indicator Output. DRHOT is an open-drain output. Pull up with an external resistor. When the device's temperature exceeds the programmed set point, DRHOT is pulled low. Internally Connected. Connect to AGND. Internal Control Circuitry Supply Input. The input voltage range is from 4.5V to 5.5V. Bypass VCC to AGND with a 1F ceramic capacitor. The maximum resistance between VCC and VDD should be 10. Phase 2 Bootstrap Flying-Capacitor Connection. An optional resistor in series with BST2 allows the DH2 pullup current to be adjusted. Phase 2 High-Side Gate-Driver Output. DH2 swings between LX2 and BST2. Phase 2 Inductor Switching Node Connection. LX2 is the internal lower supply rail for the DH2 high-side gate driver. LX2 is also the input to the skip-mode zero-crossing comparator.
4
--
TSET
5 6 7
-- -- 7
DRHOT I.C. VCC
8 9 10 11 12 13 14 15 16 17 18
8 9 10 11 12 13 14 15 16 17 18
BST2 DH2 LX2
PGND2 Phase 2 Power Ground. PGND2 is the internal lower supply rail for the DL2 low-side gate driver. DL2 VDD DL1 Phase 2 Low-Side Gate-Driver Output. DL2 swings between PGND2 and VDD. DL2 is high in shutdown. DL_ Gate-Driver Supply Input. The input voltage range is from 4.5V to 5.5V. Bypass VDD to the power ground with a 2.2F ceramic capacitor. Phase 1 Low-Side Gate-Driver Output. DL1 swings between PGND1 and VDD. DL1 is high in shutdown.
PGND1 Phase 1 Power Ground. PGND1 is the internal lower supply rail for the DL1 low-side gate driver. LX1 DH1 BST1 Phase 1 Inductor Switching Node Connection. LX1 is the internal lower supply rail for the DH1 high-side gate driver. LX1 is also the input to the skip-mode zero-crossing comparator. Phase 1 High-Side Gate-Driver Output. DH1 swings between LX1 and BST1. Phase 1 Bootstrap Flying-Capacitor Connection. An optional resistor in series with BST1 allows the DH1 pullup current to be adjusted. Pulse-Skipping-Mode Control Input. The pulse-skipping mode is enabled when SKIP is low. When SKIP is high, both drivers operate in PWM mode (i.e., except during dead times, DL_ is the complement of DH_). Shutdown Control Input. When SHDN and SKIP are low, DH_ is forced low, DL_ forced high, and the device enters into a low-power shutdown state. Temperature sensing is disabled in shutdown. No Connection. Not internally connected.
19
19
SKIP
20 --
20 4, 5, 6
SHDN N. C.
6
_______________________________________________________________________________________
Dual-Phase MOSFET Drivers with Temperature Sensor
Typical Operating Circuit
The typical operating circuit of the MAX8702 (Figure 2) shows the power-stage and gate-driver circuitry of a dualphase CPU core supply operating at 300kHz, with each phase capable of supplying 20A of load current. Table 1 lists recommended component options, and Table 2 lists the component suppliers' contact information. regulators. Each MOSFET driver is capable of driving 3nF capacitive loads with only 19ns propagation delay and 8ns typical rise and fall times. Larger capacitive loads are allowable but result in longer propagation and transition times. Adaptive dead-time control prevents shoot-through currents and maximizes converter
DBST1 +5V 2.2F VDD 10 VCC BST1 CIN1 DH1 0.22F LX1 DL1 RTSET TSET +5V 100k BST2 DRHOT SHDN DRSKP FROM CONTROLLER IC PWM1 PWM2 SKIP PWM1 PWM2 LX2 DL2 PGND2 NL2 D2 DH2 0.22F NH2 L2 VOUT COUT2 VIN 7V TO 20V CIN2 PGND1 DBST2 +5V NL1 D1 NH1 L1 VOUT COUT1 VIN 7V TO 20V
MAX8702/MAX8703
Detailed Description
The MAX8702/MAX8703 dual-phase noninverting MOSFET drivers are intended to work with PWM controller ICs in CPU core and other multiphase switching
Table 1. Component List
DESIGNATION Total Input Capacitance (CIN) Total Output Capacitance (COUT) Schottky Diode (per phase) DESCRIPTION (4) 10F, 25V Taiyo Yuden TMK432BJ106KM or TDK C4532X5R1E106M (4) 330F, 2.5V, 9m low-ESR polymer capacitor (D case) Sanyo 2R5TPE330M9 3A Schottky diode Central Semiconductor CMSH3-40 0.6H Panasonic ETQP1H0R6BFA or Sumida CDEP134H-0R6 Siliconix (1) Si7892DP or International Rectifier (2) IRF6604 Siliconix (2) Si7442DP or International Rectifier (2) IRF6603
1F AGND
MAX8702
Inductor (per phase) High-Side MOSFET (NH, per phase) Low-Side MOSFET (NL, per phase)
Figure 2. MAX8702 Typical Operating Circuit
TSET* DRHOT* VCC
TEMP SENSOR + TSDN
PWM BLOCK (x2)
BST_ DH_ LX_
Table 2. Component Suppliers
SUPPLIER Central Semiconductor Fairchild Semiconductor International Rectifier Panasonic Sanyo Siliconix (Vishay) Sumida Taiyo Yuden TDK WEBSITE www.centralsemi.com www.fairchildsemi.com www.irf.com www.panasonic.com www.secc.co.jp www.vishay.com www.sumida.com www.t-yuden.com www.component.tdk.com
AGND PWM_ SHDN
UVLO CONTROL AND ADAPTIVE DEAD-TIME CIRCUIT
MAX8702 MAX8703
VDD
LX_ PGND_ SKIP *MAX8702 ONLY ZX
DL_ PGND_
Figure 3. MAX8702 Functional Diagram _______________________________________________________________________________________ 7
Dual-Phase MOSFET Drivers with Temperature Sensor MAX8702/MAX8703
efficiency while allowing operation with a variety of MOSFETs and PWM controllers. A UVLO circuit allows proper power-on sequencing. The PWM control inputs are both TTL and CMOS compatible. The MAX8702 integrates a resistor-programmable temperature sensor. An open-drain output (DRHOT) signals to the system when the die temperature of the driver exceeds the set temperature. See the Temperature Sensor section.
CVDD VDD (RBST)* DBST INPUT (VIN)
BST
CBST DH NH L
MOSFET Gate Drivers (DH, DL)
The DH and DL drivers are optimized for driving moderately sized high-side and larger low-side power MOSFETs. This is consistent with the low duty factor seen in the notebook CPU environment, where a large VIN - VOUT differential exists. Two adaptive dead-time circuits monitor the DH and DL outputs and prevent the opposite-side FET from turning on until DH or DL is fully off. There must be a low-resistance, low-inductance path from the DH and DL drivers to the MOSFET gates for the adaptive dead-time circuits to work properly. Otherwise, the sense circuitry interprets the MOSFET gate as "off" while there is actually still charge left on the gate. Use very short, wide traces measuring 10 to 20 squares (50 to 100 mils wide if the MOSFET is 1in from the device). The internal pulldown transistor that drives DL low is robust, with a 0.35 (typ) on-resistance. This helps prevent DL from being pulled up due to capacitive coupling from the drain-to-gate capacitance of the low-side synchronous-rectifier MOSFETs when LX switches from ground to VIN. Applications with high input voltages and long, inductive DL traces may require additional gate-tosource capacitance to ensure fast-rising LX edges do not pull up the low-side MOSFET's gate voltage, causing shoot-through currents. The capacitive coupling between LX and DL created by the MOSFET's gate-todrain capacitance (CRSS), gate-to-source capacitance (CISS - CRSS), and additional board parasitics should not exceed the minimum threshold voltage: C VGS(TH) < VIN RSS CISS Lot-to-lot variation of the threshold voltage can cause problems in marginal designs. Typically, adding a 4700pF capacitor between DL and power ground, close to the low-side MOSFETs, greatly reduces coupling. To prevent excessive turn-off delays, do not exceed 22nF of total gate capacitance. Alternatively, shoot-through currents may be caused by a combination of fast high-side MOSFETs and slow low8
LX
MAX8702 MAX8703
( )* OPTIONAL--THE RESISTOR REDUCES THE SWITCHING-NODE RISE TIME.
Figure 4. High-Side Gate-Driver Boost Circuitry
side MOSFETs. If the turn-off delay time of the low-side MOSFETs is too long, the high-side MOSFETs can turn on before the low-side MOSFETs have actually turned off. Adding a resistor of less than 5 in series with BST slows down the high-side MOSFET turn-on time, eliminating the shoot-through currents without degrading the turn-off time (RBST in Figure 4). Slowing down the high-side MOSFETs also reduces the LX node rise time, thereby reducing the EMI and high-frequency coupling responsible for switching noise.
Boost Capacitor Selection
The MAX8702/MAX8703 use a bootstrap circuit to generate the floating supply voltages for the high-side drivers (DH). The boost capacitors (CBST) selected must be large enough to handle the gate-charging requirements of the high-side MOSFETs. Typically, 0.1F ceramic capacitors work well for low-power applications driving medium-sized MOSFETs. However, highcurrent applications driving large, high-side MOSFETs require boost capacitors larger than 0.1F. For these applications, select the boost capacitors to avoid discharging the capacitor more than 200mV while charging the high-side MOSFET's gates: CBST = N x QGATE 200mV
where N is the number of high-side MOSFETs used for one phase and QGATE is the total gate charge specified in the MOSFET's data sheet. For example, assume
_______________________________________________________________________________________
Dual-Phase MOSFET Drivers with Temperature Sensor
(2) IRF7811W n-channel MOSFETs are used on the high side. According to the manufacturer's data sheet, a single IRF7811W has a maximum gate charge of 24nC (VGS = 5V). Using the above equation, the required boost capacitance is: CBST = 2 x 24nC = 0.24F 200mV system power is removed without going through the proper shutdown sequence.
MAX8702/MAX8703
Low-Power Pulse Skipping
The MAX8702/MAX8703 enter into low-power pulseskipping mode when SKIP is pulled low. In skip mode, an inherent automatic switchover to pulse frequency modulation (PFM) takes place at light loads. A zerocrossing comparator truncates the low-side switch ontime at the inductor current's zero-crossing. The comparator senses the voltage across LX and PGND. Once VLX - VPGND drops below the zero-crossing comparator threshold (see the Electrical Characteristics), the comparator forces DL low. This mechanism causes the threshold between pulse-skipping PFM and nonskipping PWM operation to coincide with the boundary between continuous and discontinuous inductor-current operation. The PFM/PWM crossover occurs when the load current of each phase is equal to 1/2 the peakto-peak ripple current, which is a function of the inductor value. For a battery input range of 7V to 20V, this threshold is relatively constant, with only a minor dependence on the input voltage due to the typically low duty cycles. The switching waveforms may appear noisy and asynchronous when light loading activates the pulse-skipping operation, but this is a normal operating condition that results in high light-load efficiency.
Selecting the closest standard value, this example requires a 0.22F ceramic capacitor.
5V Bias Supply (VCC and VDD)
VDD provides the supply voltages for the low-side drivers (DL). The decoupling capacitor at V DD also charges the BST capacitors during the time period when DL is high. Therefore, the VDD capacitor should be large enough to minimize the ripple voltage during switching transitions. CVDD should be chosen according to the following equation: CVDD = 10 x CBST In the example above, a 0.22F capacitor is used for CBST, so the VDD capacitor should be 2.2F. VCC provides the supply voltage for the internal logic circuit and temperature sensor. To avoid switching noise from coupling into the sensitive internal circuit, an RC filter is recommended for the VCC pin. Place a 10 resistor from the supply voltage to the VCC pin and a 1F capacitor from the VCC pin to AGND. The total bias current IBIAS from the 5V supply can be calculated using the following equation: IBIAS = IDD + ICC IDD = nPHASE x fSW x (nNH x QG(NH) + nNL x QG(NL)) where n PHASE is the number of phases, f SW is the switching frequency, Q G(NH) and Q G(NL) are the MOSFET data sheet's total gate-charge specification limits at VGS = 5V, nNH is the total number of high-side MOSFETs in parallel, nNL is the total number of lowside MOSFETs in parallel, and ICC is the VCC supply current.
Shutdown
The MAX8702/MAX8703 feature a low-power shutdown mode that reduces the VCC quiescent current drawn to 2A (typ). Driving SHDN and SKIP low sets DH low and DL high. Temperature sensing is disabled in shutdown.
Temperature Sensor (MAX8702 Only)
The MAX8702 includes a fully integrated resistor-programmable temperature sensor. The sensor incorporates two temperature-dependent reference signals and one comparator. One signal exhibits a characteristic that is proportional to temperature, and the other is complementary to temperature. The temperature at which the two signals are equal determines the thermal trip point. When the temperature of the device exceeds the trip point, the open-drain output DRHOT pulls low.
Undervoltage Lockout (UVLO)
When VCC is below the UVLO threshold (3.85V typ) and SHDN and SKIP are low, DL is kept high and DH is held low. This provides output overvoltage protection as soon as the supply voltage is applied. Once VCC is above the UVLO threshold and SHDN is high, DL and DH levels depend on the PWM signal applied. If VCC falls below the UVLO threshold while SHDN is high, both DL and DH are immediately forced low. This prevents negative undershoots on the output when the
Table 3. Modes of Operation
SHDN L L H H SKIP L H L H MODE OF OPERATION Low-power shutdown state; temperature sensing disabled PWM operation Pulse-skipping operation PWM operation
_______________________________________________________________________________________
9
Dual-Phase MOSFET Drivers with Temperature Sensor MAX8702/MAX8703
A 10C hysteresis keeps the output from oscillating when the temperature is close to the threshold. The thermal trip point is programmable up to +160C through an external resistor between TSET and AGND. Use the following equation to determine the value of the resistor: RTSET = (85,210 / T) - (745,200 / T2) - 195 where RTSET is the value of the set-point resistor in k and T is the trip-point temperature in Kelvin. The MAX8702 and MAX8703 include a thermal-shutdown circuit that is independent of the temperature sensor. The thermal shutdown has a fixed threshold of +160C (typ) with 10C of thermal hysteresis. When the die temperature exceeds +160C, DH is pulled low and DL is pulled high. The driver automatically resets when the die temperature drops by +10C. case power dissipation due to resistance occurs at the minimum input voltage:
2 V I PD(NHRESISTIVE) = OUT LOAD RDS(ON) VIN nTOTAL
Applications Information
Power MOSFET Selection
Most of the following MOSFET guidelines focus on the challenge of obtaining high load-current capability when using high-voltage (>20V) AC adapters. Low-current applications usually require less attention. The high-side MOSFET (NH) must be able to dissipate the resistive losses plus the switching losses at both VIN(MIN) and VIN(MAX). Calculate both of these sums. Ideally, the losses at VIN(MIN) should be roughly equal to losses at VIN(MAX), with lower losses in between. If the losses at VIN(MIN) are significantly higher than the losses at VIN(MAX), consider increasing the size of NH (reducing RDS(ON) but increasing CGATE). Conversely, if the losses at VIN(MAX) are significantly higher than the losses at VIN(MIN), consider reducing the size of NH (increasing RDS(ON) but reducing CGATE). If VIN does not vary over a wide range, the minimum power dissipation occurs where the resistive losses equal the switching losses. Choose a low-side MOSFET that has the lowest possible on-resistance (RDS(ON) ), comes in a moderatesized package (i.e., one or two SO-8s, DPAK, or D2PAK), and is reasonably priced. Ensure that the DL gate driver can supply sufficient current to support the gate charge and the current injected into the parasitic gate-to-drain capacitor caused by the high-side MOSFET turning on; otherwise, cross-conduction problems can occur.
where nTOTAL is the total number of phases. Generally, a small high-side MOSFET is desired to reduce switching losses at high input voltages. However, the RDS(ON) required to stay within package power dissipation often limits how small the MOSFETs can be. Again, the optimum occurs when the switching losses equal the conduction (RDS(ON)) losses. Highside switching losses do not usually become an issue until the input is greater than approximately 15V. Calculating the power dissipation in high-side MOSFETs (NH) due to switching losses is difficult since it must allow for difficult quantifying factors that influence the turn-on and turn-off times. These factors include the internal gate resistance, gate charge, threshold voltage, source inductance, and PC board layout characteristics. The following switching-loss calculation provides only a very rough estimate and is no substitute for breadboard evaluation, preferably including verification using a thermocouple mounted on NH: C f I PD(NHSWITCHING) = (VIN(MAX))2 RSS SW LOAD IGATE nTOTAL where CRSS is the reverse transfer capacitance of NH and IGATE is the peak gate-drive source/sink current (5A typ). Switching losses in the high-side MOSFET can become an insidious heat problem when maximum AC adapter voltages are applied, due to the squared term in the C x VIN2 x fSW switching-loss equation. If the high-side MOSFET chosen for adequate RDS(ON) at low battery voltages becomes extraordinarily hot when biased from V IN(MAX) , consider choosing another MOSFET with lower parasitic capacitance. For the low-side MOSFET (NL), the worst-case power dissipation always occurs at the maximum input voltage: V PD(NLRESISTIVE) = 1- OUT VIN(MAX)
2 ILOAD RDS(ON) nTOTAL
MOSFET Power Dissipation
Worst-case conduction losses occur at the duty factor extremes. For the high-side MOSFET (NH), the worst-
The worst case for MOSFET power dissipation occurs under heavy overloads that are greater than ILOAD(MAX) but are not quite high enough to exceed
10
______________________________________________________________________________________
Dual-Phase MOSFET Drivers with Temperature Sensor
the current limit and cause the fault latch to trip. The MOSFETs must have a good-sized heatsink to handle the overload power dissipation. The heat sink can be a large copper field on the PC board or an externally mounted device. The Schottky diode only conducts during the dead time when both the high-side and low-side MOSFETs are off. Choose a Schottky diode with a forward voltage low enough to prevent the low-side MOSFET body diode from turning on during the dead time, and a peak current rating higher than the peak inductor current. The Schottky diode must be rated to handle the average power dissipation per switching cycle. This diode is optional and can be removed if efficiency is not critical. 2) Minimize the high-current loops from the input capacitor, upper-switching MOSFET, and low-side MOSFET back to the input capacitor negative terminal. 3) Provide enough copper area at and around the switching MOSFETs and inductors to aid in thermal dissipation. 4) Connect the PGND1 and PGND2 pins as close as possible to the source of the low-side MOSFETs. 5) Keep LX traces away from sensitive analog components and nodes. Place the IC and analog components on the opposite side of the board from the power-switching node if possible. 6) Use two or more vias for DL and DH traces when changing layers to reduce via inductance. Figure 5 shows a PC board layout example.
MAX8702/MAX8703
IC Power Dissipation and Thermal Considerations
Power dissipation in the IC package comes mainly from driving the MOSFETs. Therefore, it is a function of both switching frequency and the total gate charge of the selected MOSFETs. The total power dissipation when both drivers are switching is given by: PD(IC) = IBIAS x 5V where IBIAS is the bias current of the 5V supply calculated in the 5V Bias Supply (VDD and VCC) section . The rise in die temperature due to self-heating is given by the following formula: TJ = PD(IC) x JA where PD(IC) is the power dissipated by the device, and JA is the package's thermal resistance. The typical thermal resistance is 59.3C/W for the 4mm x 4mm thin QFN package. For example, if the MAX8702 dissipates 500mW of power within the IC, this corresponds to a 30C shift in the die temperature in the thin QFN package.
VIA TO POWER GROUND USE DOUBLE VIAS FOR DL_
CONNECT AGND AND PGND_ BENEATH THE CONTROLLER AT ONE POINT ONLY AS SHOWN
INPUT CIN CIN POWER GROUND CIN CIN
PC Board Layout Considerations
The MAX8702/MAX8703 MOSFET drivers source and sink large currents to drive MOSFETs at high switching speeds. The high di/dt can cause unacceptable ringing if the trace lengths and impedances are not well controlled. The following PC board layout guidelines are recommended when designing with the device: 1) Place VCC and VDD decoupling capacitors as close to their respective pins as possible.
INDUCTOR INDUCTOR
COUT
Figure 5. PC Board Layout Example
______________________________________________________________________________________
COUT
OUTPUT
COUT
COUT
11
Dual-Phase MOSFET Drivers with Temperature Sensor MAX8702/MAX8703
Pin Configuration
SHDN DH1 LX1
Chip Information
TRANSISTOR COUNT: 1100 PROCESS: BiCMOS
TOP VIEW
SKIP
BST1
19
PWM1 PWM2 AGND TSET* DRHOT*
1 2 3 4 5 10 7 6 8 9
20
18
17
16 15 14
PGND1 DL1 VDD DL2 PGND2
MAX8702 MAX8703
13 12 11
BST2
I.C.*
VCC
THIN QFN (4mm x 4mm) *THESE PINS ARE N.C. ON THE MAX8703
12
______________________________________________________________________________________
DH2
LX2
Dual-Phase MOSFET Drivers with Temperature Sensor
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.)
MAX8702/MAX8703
PACKAGE OUTLINE 12, 16, 20, 24L THIN QFN, 4x4x0.8mm
21-0139
C
1 2
______________________________________________________________________________________
13
24L QFN THIN.EPS
Dual-Phase MOSFET Drivers with Temperature Sensor MAX8702/MAX8703
Package Information (continued)
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.)
PACKAGE OUTLINE 12, 16, 20, 24L THIN QFN, 4x4x0.8mm
21-0139
C
2 2
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
14 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2004 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

▲Up To Search▲

Price & Availability of MAX8702

	To Download MAX8702 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .